Method and system for multi-carrier time division multiplexing modulation/demodulation

ABSTRACT

The present disclosure provides a multi-carrier time-division multiplexing (MC-TDMA) modulation and demodulation method and system. Before multi-carrier modulation is performed on an input symbol, an interleaving allocation and an FFT may be performed, a time domain symbol may be transformed into a frequency domain symbol signal to perform a MDFT treatment. A sending end may adopt an analyzing filter bank structure, and pre-filtering and an IFFT may be performed on a signal successively. A pre-filter may be positioned between an NM point FFT and an M point IFFT, a PAPR value of the system may be reduced using the symmetry of a coefficient of a filter, and a frequency domain symbol signal may be allocated to different sub-bands for multi-carrier modulation.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. Ser. No. 16/683,411 filed onNov. 14, 2019, which is a continuation of U.S. Ser. No. 15/563,574 (nowU.S. Pat. No. 10,541,846) filed on Sep. 30, 2017, which is a U.S.national stage of International Application No. PCT/CN2015/075557, filedon Mar. 31, 2015, the contents of which are incorporated herein byreference in its entirety.

TECHNICAL FIELD

The present disclosure relates to multi-carrier modulation anddemodulation technologies, and particularly to a multi-carriertime-division multiplexing (MC-TDMA) modulation/demodulation method andsystem.

BACKGROUND

The mobile communication technology has undergone development of fourgenerations. The first-generation wireless communication belongs toanalog communication, and only voice signals can be transmitted. Fromthe second-generation wireless network, a mobile communication networkcomes into a digital network era, and voice and data can be transmittedsimultaneously. With the development of a modulation technology, a datatransfer rate is increased from 14.4 Kbps to 1 Gbps from thesecond-generation to fourth-generation mobile communication (Long TermEvolution (LTE)). The second-generation wireless communication adoptsthe Gaussian minimum shift keying (GMSK) modulation technology, thethird-generation mobile communication adopts quadrature phase shiftkeying (QPSK), and both of the modulation technologies belong to thesingle-carrier modulation.

In order to increase the data transfer rate, the fourth-generationmobile communication network typified by the LTE standard adopts amulti-carrier modulation technology. In the LTE standard, thesingle-carrier frequency-division multiple access (SC-FDMA) technologymay be adopted in the uplink, and the orthogonal frequency-divisionmultiple access (OFDMA) technology may be adopted in the downlink. Inthe multi-carrier modulation, a high-speed signal may be divided into aplurality of low-speed signals through an IFFT, the low-speed signalsmay be modulated into different sub-carriers, and a signal having a longsymbol period may be synthesized for transmission. Therefore, themulti-carrier modulation technology may have a capability ofanti-multipath fading in a wireless channel.

A major impetus for the development of future mobile communication maybe Internet of Things (IoT) and machine-to-machine (M2M) communication,and characteristics of the IoT and M2M communication may be random,asynchrony, short data, low latency, low power consumption and low cost.Therefore, there may be a need to study a new modulation technology tomeet the requirements of the development of the future wirelesscommunication, and thereby the present disclosure is provided.

SUMMARY

A multi-carrier time-division multiplexing modulation method, whereinthe characteristics may lie in performing an interleaving allocation onan input symbol; performing a fast Fourier transform (FFT) using a fastFourier transform; transforming a time domain symbol into a frequencydomain symbol signal to perform a modified discrete Fourier transform(MDFT) treatment.

According to another embodiment of the present disclosure, the MDFTtreatment in the multi-carrier time-division multiplexing modulationmethod may further include a sub-band analyzing and filtering treatment,an inverse Fourier transform treatment, and an interleaving operationtreatment.

According to another embodiment of the present disclosure, the fastFourier transform in the multi-carrier time-division multiplexingmodulation method may be further an NM-point fast Fourier transform, Nand M herein being positive integers that are larger than or equal to 1.

According to another embodiment of the present disclosure, the sub-bandanalyzing and filtering treatment included in the MDFT treatment in themulti-carrier time-division multiplexing modulation method may furtherinclude performing pre-filtering on an NM point frequency domain symbolsignal, constructing a coefficient matrix H based on a prototype filterfunction, and right-multiplying the NM point frequency domain symbolsignal by the coefficient matrix H to generate a 2NM point frequencydomain symbol signal.

According to another embodiment of the present disclosure, thecoefficient matrix used in the multi-carrier time-division multiplexingmodulation method may be further generated based on performing an M/2circular right-shift on a matrix with 4N×2N matrix element blocks.

According to another embodiment of the present disclosure, thecoefficient matrix used in the multi-carrier time-division multiplexingmodulation method may further include the following sub-matrices hi,0and hi,1: dividing a coefficient h(n) (0<=n<=NM−1) of a root raisedcosine (RRC) prototype function into N sub-blocks (each of the Nsub-blocks includes M points), and constructing the diagonal matrix hi,0and the diagonal matrix hi,1 based on the first M/2 points and the lastM/2 points of the ith sub-block of the N sub-blocks respectively, iherein is an integer between 0 and N−1.

According to another embodiment of the present disclosure, anarrangement of the sub-matrix hi,0 and the sub-matrix hi,1 included inthe coefficient matrix H used in the multi-carrier time-divisionmultiplexing modulation method may be as follows:

$H = \begin{bmatrix}h_{0,0} & 0 & h_{1,0} & 0 & h_{2,0} & \cdots & 0 & h_{{N - 1},0} & 0 \\0 & h_{0,1} & 0 & h_{1,1} & 0 & \cdots & h_{{N - 2},1} & 0 & h_{{N - 1},1} \\0 & h_{0,0} & 0 & h_{1,0} & 0 & \cdots & h_{{N - 2},0} & 0 & h_{{N - 1},0} \\h_{{N - 1},1} & 0 & h_{0,1} & 0 & h_{1,1} & \cdots & 0 & h_{{N - 2},1} & 0 \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots \\h_{1,0} & 0 & h_{2,0} & 0 & h_{3,0} & \cdots & 0 & h_{0,0} & 0 \\0 & h_{1,1} & 0 & h_{2,1} & 0 & \cdots & h_{{N - 1},1} & 0 & h_{0,1}\end{bmatrix}$

According to another embodiment of the present disclosure, the inverseFourier transform included in the MDFT treatment in the multi-carriertime-division multiplexing modulation method may be further an M-pointinverse Fourier transform repeated 2N times.

The present disclosure also provides a multi-carrier time-divisionmultiplexing demodulation method, and the method may include: removingprefixes of a received signal, equalizing the received signal withoutthe prefixes by an equalizer, performing an inverse modified discreteFourier transform (IMDFT) treatment to synthesize a full-band frequencydomain signal, performing an inverse fast Fourier transform (IFFT) onthe synthesized frequency domain signal to generate a time domainsignal, performing symbol inverse ordering on the generated time domainsignal, and obtaining a reconstructed input symbol signal of a sendingend through a symbol inverse mapping treatment.

According to another embodiment of the present disclosure, the IMDFTtreatment in the multi-carrier time-division multiplexing demodulationmethod may further include an inverse interleaving operation treatment,a Fourier transform treatment, and a sub-band synthesizing and filteringtreatment.

According to another embodiment of the present disclosure, the inverseFourier transform in the multi-carrier time-division multiplexingdemodulation method may be further an NM-point inverse fast Fouriertransform.

According to another embodiment of the present disclosure, the Fouriertransform included in the IMDFT treatment in the multi-carriertime-division multiplexing demodulation method may be further an M-pointFourier transform repeated 2N times.

According to another embodiment of the present disclosure, the sub-bandsynthesizing and filtering treatment included in the IMDFT treatment inthe multi-carrier time-division multiplexing demodulation method mayfurther include performing post-filtering on a 2NM point frequencydomain symbol signal, and right-multiplying the 2NM point frequencydomain symbol signal by a transpose matrix of a coefficient matrix H togenerate an NM point frequency domain symbol signal.

A multi-carrier time-division multiplexing modulation system mayinclude: a symbol mapping unit, a symbol ordering unit, a unitconfigured to implement a fast Fourier transform, and an MDFT unit; thesaid MDFT unit may include a sub-band analyzing and filtering module, aninverse Fourier transform module, and an interleaving operation module.

According to another embodiment of the present disclosure, the Fouriertransform implemented by the unit configured to implement the fastFourier transform included in the multi-carrier time-divisionmultiplexing modulation system may be further an NM-point fast Fouriertransform, N and M being positive integers that are larger than or equalto 1.

According to another embodiment of the present disclosure, the sub-bandanalyzing and filtering module included in the MDFT unit in themulti-carrier time-division multiplexing modulation system may befurther used to perform pre-filtering on an NM point frequency domainsymbol signal, construct a coefficient matrix H based on a prototypefilter function, and right-multiply the NM point frequency domain symbolsignal by the coefficient matrix H to generate a 2NM point frequencydomain symbol signal.

According to another embodiment of the present disclosure, the inverseFourier transform implemented by the inverse Fourier transform moduleincluded in the MDFT unit in the multi-carrier time-divisionmultiplexing modulation system may be further an M-point inverse Fouriertransform repeated 2N times.

A multi-carrier time-division multiplexing demodulation system mayinclude an IMDFT unit, a unit configured to implement an inverse fastFourier transform, a symbol inverse ordering unit, a symbol inversemapping unit; the said IMDFT unit may include an inverse interleavingoperation module, a Fourier transform module, and a sub-bandsynthesizing and filtering module.

According to another embodiment of the present disclosure, the inversefast Fourier transform implemented by the unit configured to implementthe inverse fast Fourier transform included in the multi-carriertime-division multiplexing demodulation system may be further anNM-point inverse fast Fourier transform, N and M being positive integersthat are larger than or equal to 1.

According to another embodiment of the present disclosure, the sub-bandsynthesizing and filtering module included in the IMDFT unit in themulti-carrier time-division multiplexing demodulation system may befurther used to perform post-filtering on a 2NM point frequency domainsymbol signal, and right-multiply the 2NM point frequency domain symbolsignal by a transpose matrix of a coefficient matrix H to generate an NMpoint frequency domain symbol signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a structural block diagram illustrating a principle of aMC-TDMA modulation system;

FIG. 2 is a block diagram illustrating a separation implementation of afrequency domain MDFT filter bank in MC-TDMA;

FIG. 3 is a schematic diagram of a PAPR simulation result;

FIG. 4 illustrates a simulation result of a symbol error rate of aMC-TDMA system; and

FIG. 5 illustrates a simulation result of an anti-carrier drift of aMC-TDMA system.

DETAILED DESCRIPTION OF EMBODIMENTS

As used in the disclosure and the appended claims, the singular forms“a,” “an,” and “the” include plural referents unless the content clearlydictates otherwise. In general, the terms “comprises,” “comprising,”“includes,” and/or “including” when used in the disclosure, specify thepresence of stated steps and elements, but do not preclude the presenceor addition of one or more other steps or elements.

A multi-carrier modulation technology may be widely used in thefourth-generation mobile communication network typified by the LTEstandard, and particularly, the orthogonal frequency division multipleaccess (OFDMA) technology may be adopted in the downlink. In themulti-carrier modulation, a high-speed signal may be divided into aplurality of low-speed signals through an inverse fast Fourier transform(IFFT), the low-speed signals may be modulated into differentsub-carriers, and a signal having a long symbol period may besynthesized for transmission. Therefore, the multi-carrier modulationtechnology has a natural capability of anti-multipath fading in awireless channel. Since the symbol period is expanded, the capacity of asignal to resist the multipath fading may be improved greatly, and thusthe multi-carrier modulation technology may be an indispensable portionof high-speed wireless communication.

OFDMA and SC-FDMA may be sensitive to a carrier drift. In addition, apeak to average power ratio (PAPR) value of OFDMA may be relativelylarge. In contrast, a PAPR value of SC-FDMA may be relatively small. TheLTE standard may be adopted in the present fourth-generation mobilecommunication network, for example, SC-FDMA is adopted in uplinkcommunication. Since there is a need to remain multi-carriercharacteristics of a SC-FDMA modulation signal, localized SC-FDMA(SC-LFDMA) may be adopted in the LTE standard. Although a PAPR value ofSC-LFDMA is less than that of OFDMA, the PAPR value of SC-LFDMA may bestill very different from a theoretical minimum value of PAPR, forexample, it is possible to improve the PAPR of SC-LFDMA. Sub-carrierfrequencies of a sending end and a receiving end may need to maintainstrict synchronization in OFDMA and SC-FDMA, and thus there areextremely high requirements for crystal oscillator accuracies of thereceiving end and the sending end.

The reason for the large PAPR value of the present multi-carriermodulation system may be that an inverse fast Fourier transform (IFFT)is used. Since a basis function of an IFFT is a complex exponentialfunction, an amplitude value of the complex exponential function may beincreased by multiplying and superimposing random symbols. Especially,when phases of the complex exponential are consistent after multiplying,the amplitude value may be maximum while the PAPR value is also maximum.There are many methods for reducing the PAPR value in the OFDMA system.However, these methods may only solve the problem of the high PAPRvalue, and cannot completely address the effect of sub-carrier drift ofthe OFDMA system on the system performance.

An alternative method for improving the anti-carrier drift capability ofthe multi-carrier modulation system is to replace the IFFT in the OFDMAsystem with a filter bank. A frequency characteristic of a filter bankprototype function may be better than that of a rectangular windowfunction in the IFFT, and therefore inter-subcarrier interference (ICI)can be well cancelled. In addition, the good frequency characteristic ofthe prototype function may also increase the power spectral density ofthe system, reduce power leak between sub-bands, and improve thevalidity of signal transmission. However, a filter bank multi-carrier(FBMC) modulation system may have two features of high PAPR value andlong delay since the good frequency characteristic requires a largeprototype filter function coefficient. These features may restrict theapplication of the FBMC modulation system in practice.

An analysis may show that there are two basic methods to reduce the PAPRvalue, one is to reduce peak power of a modulation symbol, and the otheris to shorten the length of the IFFT. Shortening the length of the IFFTwhile reducing frequency resolution of the modulation system may improvethe anti-carrier drift capability of the system. Since the theoreticalminimum PAPR value is a PAPR of an original input symbol signal, thePAPR value is lower when an output symbol is modulated as close aspossible to the original input symbol. In the single carrier modulation,an FFT carrier mapping may be firstly performed on an input symbol, andthen IFFT modulation may be performed. Based on the mode of a carriermapping, a modulation symbol after FFT/IFFT may be the weighted sum(localized mapping) of the input symbol or repeat (interleaving mapping)of the input symbol. The PAPR value may directly influence the batterylifetime of a mobile terminal, and therefore, the lower PAPR value maybe better for uplink communication.

At present, all modulation methods and systems (including systems whichhave been used in the standard) cannot solve the problems of the highPAPR in OFDMA and high carrier drift sensitivity at the same time. Thepresent disclosure may combine the advantages of SC-FDMA and FBMCmodulation technologies, and in a modulation system, successfully solvethe above two problems which have plagued the wireless communicationphysics layer for many years. The MC-TDMA modulation system provided bythe present disclosure can be used not only for high-speed communicationbut also for IoT and M2M communication.

A specific embodiment of the present disclosure may be a multi-carriertime-division multiplexing modulation and demodulation method, whereinbefore multi-carrier modulation is performed on an input symbol, aninterleaving allocation and an FFT may be performed on the input symbol,a time domain symbol may be transformed into a frequency domain symbolsignal, and an MDFT treatment may be performed on the frequency domainsymbol signal. It should be noted that the description herein is merelya major process of a specific embodiment and should not be regarded as aunique embodiment, the various steps therein may be not necessary, andthe whole process and specific steps thereof are not limited to thedescription of the drawings and the above description. For example,before an interleaving allocation is performed on an input symbol, somecorresponding preprocessing steps may be carried out, and the order ofan FFT and an MDFT treatment may be adjusted appropriately. It will beapparent to those skilled in the art that after understanding thecontents and principles of the present disclosure, various modificationsand variations can be made in the form and details of the system withoutdeparting from the principles and constructions of the presentdisclosure. However, these modifications and variations are still withinthe scope of the claims of the present disclosure.

In the portion of the MDFT treatment, the difference between the presentdisclosure and the conventional FBMC (OFDM/OQAM) modulation scheme andthe classical MDFT treatment structure may be that a sending end adoptsa synthesizing filter bank structure in the traditional FMBC, but in thepresent disclosure, a sending end adopts an analyzing filter bankstructure, and an IFFT is performed instead of the conventional FFTafter pre-filtering is performed on a signal. A pre-filter may bepositioned between an NM-point FFT and an M-point IFFT, and in MC-TDMA,play a dual role of reducing the PAPR value of the system using thesymmetry of a filter coefficient and allocating a frequency domainsymbol signal to different sub-bands for multi-carrier modulation. Thedescription herein is merely a major process of a specific embodimentand should not be regarded as a unique embodiment, the various stepstherein may be not necessary, and the whole process and specific stepsthereof are not limited to the description of the drawings and the abovedescription.

A further specific embodiment of the present disclosure may provide amulti-carrier time division multiplexing system, and the system mayinclude a sending end and a receiving end. The sending end may include:a symbol mapping unit for performing a symbol mapping on a binary bitsequence to obtain a complex symbol signal; a symbol ordering unit forperforming an interleaving allocation on an input symbol; a fast Fouriertransform (FFT) unit for transforming a time domain symbol signal into afrequency domain symbol signal; and an MDFT unit, which adopts ananalyzing filter bank structure, for performing an IFFT after performingpre-filtering on the frequency domain symbol signal as a pre-filter. Thereceiving end may include: a prefix removal unit and an equalizer forrespectively performing prefix removal and an equalization treatment ona received signal to obtain a symbol signal; an IMDFT unit forperforming an inverse interleaving operation and performing an inverseMDFT on the signal obtained by the inverse interleaving operation; andan IFFT unit for performing an inverse fast Fourier transform (IFFT) onthe obtained symbol signal to obtain a reconstructed signal of a sendingend. The description herein is merely a major process of a specificembodiment and should not be regarded as a unique embodiment, thevarious steps therein may be not necessary, and the whole process andspecific steps thereof are not limited to the description of thedrawings and the above description. For example, depending on thecommunication quality of a wireless channel, the equalizer herein mayadopt a frequency domain zero forcing equalizer or a non-zero forcingequalizer, adopt a blind equalizer or a non-blind equalizer, or adopt anadaptive equalizer or a non-adaptive equalizer. Similarly, it will beapparent to those skilled in the art that after understanding thecontents and principles of the present disclosure, various modificationsand variations can be made in the form and details of the system withoutdeparting from the principles and constructions of the presentdisclosure. However, these modifications and variations are still withinthe scope of the claims of the present disclosure.

Further, a pre-filter of a sending end may be positioned between an NMpoint FFT and an M point IFFT, a coefficient of the filter may have thesymmetry, and a frequency domain symbol signal may be allocated todifferent sub-bands for multi-carrier modulation.

In a system, in which the number of access users is N, the number ofsub-carriers of each user is M, and thus the total number ofsub-carriers L is equal to NM. The symbol mapping unit may perform asymbol mapping to obtain M time domain input symbol signals, the symbolordering unit may allocate the M time domain input symbol signals to NMtime points using an interleaving allocation mode, and the FFT unit mayperform an FFT operation on an NM point time domain symbol signal toobtain an NM point frequency domain symbol signal.

Further, the MDFT unit may include: a sub-band analyzing and filteringportion, which is used to perform an FFT on an NM point input symbolsignal of a sending end to obtain an NM point frequency domain symbolsignal, construct a coefficient matrix H based on a prototype filterfunction, and right-multiply the NM point frequency domain symbol signalby the coefficient matrix H to obtain a 2NM point frequency domainsymbol signal; an IFFT transform portion, which is used to perform anM-point IFFT repeated 2N times on the 2NM point frequency domain symbolsignal to obtain a 2NM point time domain signal; and an interleavingoperation portion, which is used to perform an interleaving operation onthe 2NM point time domain signal to obtain an NM point output complexsymbol signal. The interleaving operation portion may divide the 2NMpoint time domain complex signal into an upper path and a lower path.Signals in the upper path may be not delayed, and there may be one bitdelay in signals in the lower path. After two times sampling isperformed on the signals in the upper and lower paths, a real part andan imaginary part may be extracted alternately to synthesize a complexsymbol signal, which makes the 2NM point complex signal become an NMpoint complex signal. The coefficient matrix H may be obtained byperforming an M/2 circular right-shift on a coefficient of an MDFTanalyzing filter bank. The coefficient of the MDFT analyzing filter bankmay be constituted by a square root raised cosine (RRC) function, andthe dimension of the matrix H may be 2NM×NM. The said constructing thecoefficient matrix H may specifically include: obtaining a matrix H of2NM×NM by performing an M/2 circular right-shift on a matrix with M×NMmatrix element blocks, and shifting M/2 points shifted from the right toM/2 points at the left. The shift may begin from the first M×NM matrixblock to the (2N−1)th matrix block.

Further, an equalizer of the receiving end may perform prefix removal ona symbol signal and then perform an FFT to obtain a frequency domainsignal, divide the frequency domain signal by a system function of achannel, and obtain a signal with channel interference removed after anIFFT. An IMDFT unit of the receiving end may be used to reproduce the NMpoint symbol signal with the channel interfere removed to a 2NM pointsymbol signal, perform an M-point FFT repeated 2N times on the 2NM pointsymbol signal to obtain a 2NM point frequency domain signal, andright-multiply the 2NM point frequency domain signal by a transposedmatrix H of the coefficient matrix H to obtain an NM point signal.

The MC-TDMA system provided by the present disclosure may be implementedby a simple structure, the sending end may be implemented by an inputsymbol interleaving allocation and an MDFT filter bank, and thereceiving end may adopt an IMDFT synthesizing filter bank structure andan FFT unit. The interleaving operation may cancel interference betweenadjacent sub-bands, and a 2NM point symbol signal is transformed to anNM point symbol signal after the interleaving operation. The number ofsub-carriers may be M, N multi-carrier modulation symbols may berespectively transmitted in N time periods, and the optimal timefrequency resolution of the system may be obtained by adjusting N and M,so that the system has a capability of anti-multipath fading in awireless channel and an anti-carrier drift capability, and the MC-TDMAsystem focus advantages of all other modulation systems. The sending endmay adopt an analyzing filter bank structure. The receiving end mayadopt a synthesizing filter bank structure and recover an NM point inputsignal of the sending end one time, so that the system has theadvantages of short time delay, saving the resource occupied by theprefix, using the frequency domain zero forcing equalizer and easyimplementation.

The implementation of the present disclosure will be described infurther detail with reference to the drawings and specific examples.Obviously, drawings described below are only some embodiments of thepresent disclosure. Those having ordinary skills in the art, withoutfurther creative efforts, may apply the present disclosure to othersimilar scenarios according to these drawings. Unless stated otherwiseor obvious from the context, the same reference numeral in the drawingsrefers to the same structure and operation.

FIG. 1 a structural block diagram illustrating a principle of a MC-TDMAmodulation system. The MC-TDMA modulation system may include a sendingend and a receiving end. The sending end may include a symbol mappingunit, a symbol ordering unit, an NM-point FFT unit, a sub-band analyzingand filtering unit, an M-point IFFT repeated 2N times unit and aninterleaving operation unit. The receiving end may include an inverseinterleaving operation unit, an M-point FFT repeated 2N times unit, asub-band synthesizing filter, an NM-point IFFT unit, a symbol inverseordering unit and a symbol inverse mapping unit.

An input binary bit sequence s(n) may be subjected to a symbol mappingby the symbol mapping unit, a QPSK or QAM method may be used in thesymbol mapping, and a complex symbol signal to be modulated may beobtained after the symbol mapping. The complex symbol signal may besubjected to a symbol ordering treatment by the symbol ordering unit,which adopts an interleaving ordering mode, for example, a symbol signalis inserted every N points, a value between two non-zero symbol signalsis zero for a single user (uplink transmission), and a value between twonon-zero symbol signals is a signal of other users. After treatment ofthe ordering unit, an NM point symbol signal may be obtained. Theobtained ordering symbol signal may be processed by the NM-point FFTunit, after the sub-band analyzing and filtering unit performs analyzingand filtering on the signal, the signal may be transmitted to theM-point IFFT repeated 2N times unit for transform to obtain a signalafter modulation, and the obtained signal may be transmitted to achannel after an interleaving operation and prefixing and then sent tothe receiving end. The receiving end may remove prefixes of the receivedsignal, and the equalizer may perform equalization processing on thesignal without the prefixes. After the signal is transmitted to theinverse interleaving operation unit for processing, the M-point FFT unitmay perform an M-point FFT repeated 2N times on the signal to obtain afrequency domain sub-band signal. Then, a full-band signal may besynthesized through sub-band synthesizing and filtering, the synthesizedfrequency domain signal may be transmitted to the NM-point IFFT unit toobtain a time domain signal. The signal output by the IFFT unit may besubjected to a symbol inverse ordering, and a reconstructed input symbolsignal of the sending end may be obtained after processing of the symbolinverse mapping unit.

Specifically, for a multi-user access modulation system, it may beassumed that the total number of sub-carriers L is equal to NM, and thenumber of sub-carriers allocated to each user is M. M symbol signals maybe obtained by mapping the sub-carrier signal via the symbol mappingunit, and allocated to L time points via the interleaving operation unitadopting an interleaving symbol allocation mode. An L point FFT may beperformed on a complex symbol signal, and a time domain symbol signalmay be transformed to frequency domain to obtain a frequency domainsignal.

As shown in FIG. 2, an MDFT filter bank may be at the front of an FFTunit and composed of three portions. The first portion may be apre-filtering unit, the second portion may be an IFFT unit, and thethird portion may be a signal interleaving operation unit.

The MDFT filter bank may include: a sub-band analyzing and filteringportion, an IFFT portion, a signal interleaving operation portion. Afterthe sub-band analyzing and filtering portion and the IFFT portionrespectively completing pre-filtering and an IFFT, the signalinterleaving operation portion may perform an interleaving operation ona 2NM point symbol signal and obtain an NM point complex symbol tooutput for transmission.

The pre-filtering unit may be composed of an analyzing filter bank witha decimation ratio of M/2 (the number of channels of the filter bank isM), and perform a sub-band analyzing treatment on an input signal x(n)to obtain M sub-band signals. A filter bank with a coefficient h(n) of aprototype function may be selected, and z⁻¹ in FIG. 2 may represent onebit delay. The NM point input symbol signal x(n) of each frame may enterthe pre-filter along a delay line, subjected to M/2 sampling, obtain NMpath output, and obtain M path output after superimposing the NM pathoutput. Wherein N is determined by the number of users of the modulationsystem, and M is determined by the number of sub-carriers. After inversefast Fourier transform (IFFT) is performed on the M path output, an Mpath modulation signal may be obtained. An interleaving operation may beperformed on the modulation signal before being sent, thereby cancelinginterference between sub-bands. The interleaving operation may includethe interleaving operation of the sending end and the inverseinterleaving operation of the receiving end. The interleaving operationportion of the sending end may be composed of a decimator with adecimation value of two, real part and imaginary part operation units ofa complex number. A signal may be divided into an upper path and a lowerpath after entering the interleaving operation portion, there may be nodelay in the upper path and there may be one bit delay in the lowerpath. After two times sampling is performed on the signals in the upperand lower paths, the real part and imaginary part may be extractedalternately, and extracted real part and imaginary part values maysynthesize a new complex symbol signal value for transmission. Real partand imaginary part operations of adjacent two channels may needinterleaving, and if the real part operation is performed on an upperpath of a previous channel, the imaginary part operation may need to beperformed on an upper path of a current channel. In FIG. 2, z⁻¹ mayrepresent one bit delay, ↓ M/2 may represent M/2 sampling, ↑ M/2 mayrepresent M/2 interpolation, h(n) may represent a RRC prototype filterfunction, Re{ } and IM{ } may respectively represent the real partoperation and the imaginary part operation, x(n) and {circumflex over(x)}(n) may respectively represent an input signal of the sending endand a reconstructed signal of the receiving end.

Since the two times sampling and the interleaving operation, a 2NM pointinput may become an NM point output, thereby guaranteeing the conformityof the number of input symbols and the number of output in the MC-TDMAsystem. After the above processing, a cyclic prefix may be added on thesynthesized complex symbol signal, and then the signal may enter awireless channel to be sent.

The optimal MDFT filter bank may be obtained in the following manner.The pre-filtering portion may construct a coefficient matrix H based ona prototype filter function, and right-multiply an NM point frequencydomain symbol signal by the coefficient matrix H to obtain a 2NM pointfrequency domain symbol signal. The IFFT portion may perform an M-pointIFFT repeated 2N times on a 2NM point frequency domain symbol signal toobtain a 2NM point time domain signal. The interleaving operationportion may perform an interleaving operation on the 2NM point timedomain signal to obtain an NM point complex symbol signal output.

Specifically, the method for constructing the coefficient matrix H basedon the prototype filter function may be as follows. If the prototypefilter is a square root raised cosine (RRC) function, the coefficienth(n) of the prototype filter function may be expressed as:

${{h(n)} = \frac{{\frac{4{rn}}{M}{\cos\left\lbrack \frac{{\pi\left( {1 + r} \right)}n}{M} \right\rbrack}} + {\sin\left\lbrack \frac{{\pi\left( {1 - r} \right)}n}{M} \right\rbrack}}{\left\lbrack {1 - \left( \frac{4{rn}}{M} \right)^{2}} \right\rbrack\pi\; n}},{{- \infty} \leq n < \infty}$${h(0)} = {\frac{1}{M} + {\frac{r}{M}\left( {\frac{4}{\pi} - 1} \right)}}$${h\left( {\pm \frac{M}{4r}} \right)} = {{- \frac{r}{M}}\left\{ {{\frac{2}{\pi}{\cos\left\lbrack {\frac{\pi}{4r}\left( {1 + r} \right)} \right\rbrack}} - {\cos\left\lbrack {\frac{\pi}{4r}\left( {1 - r} \right)} \right\rbrack}} \right\}}$

wherein M is equal to the number of sub-carriers, r represents aroll-off factor of the RRC function and determines a stop bandattenuation factor of a RRC function filter, and a range of a variant nof the RRC function is determined by the length NM of the pre-filter.The construction of the coefficient matrix H is not limited to the abovemethod and the specific set parameters.

For constructing the coefficient matrix H, it may be assumed that thenumber of access users is N, the number of sub-carriers allocated toeach user is M, and the matrix H may be obtained by performing an M/2circular right-shift on a matrix with 4N×2N matrix element blocks,wherein the matrix element blocks hi,0 and hi,1 (0<=i<=N−1) may be twodiagonal matrixes. If the coefficient h(n) of the RRC prototype functionis divided into N sub-blocks (each sub-block includes M points), hi,0and hi,1 may be respectively composed of the first M/2 points and thelast M/2 points of the ith sub-block. The size of H may be 2NM×NM. Inthe process of circular shift, the M/2 points shifted from the right maybe shifted to the left M/2 points. The shift may begin from the firstM×NM matrix block to the (2N−1)th matrix block.

$H = \begin{bmatrix}h_{0,0} & 0 & h_{1,0} & 0 & h_{2,0} & \cdots & 0 & h_{{N - 1},0} & 0 \\0 & h_{0,1} & 0 & h_{1,1} & 0 & \cdots & h_{{N - 2},1} & 0 & h_{{N - 1},1} \\0 & h_{0,0} & 0 & h_{1,0} & 0 & \cdots & h_{{N - 2},0} & 0 & h_{{N - 1},0} \\h_{{N - 1},1} & 0 & h_{0,1} & 0 & h_{1,1} & \cdots & 0 & h_{{N - 2},1} & 0 \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots \\h_{1,0} & 0 & h_{2,0} & 0 & h_{3,0} & \cdots & 0 & h_{0,0} & 0 \\0 & h_{1,1} & 0 & h_{2,1} & 0 & \cdots & h_{{N - 1},1} & 0 & h_{0,1}\end{bmatrix}$ ${h_{i,0} = \begin{bmatrix}{h({iM})} & 0 & \cdots & 0 \\0 & {h\left( {{iM} + 1} \right)} & \cdots & 0 \\\vdots & \vdots & \ddots & \vdots \\0 & 0 & \cdots & {h\left( {{iM} + {M\text{/}2} + 1} \right)}\end{bmatrix}},{h_{i,1} = \begin{bmatrix}{h\left( {{iM} + {M\text{/}2}} \right)} & 0 & \cdots & 0 \\0 & {h\left( {{iM} + {M\text{/}2} + 1} \right)} & \cdots & 0 \\\vdots & \vdots & \ddots & \vdots \\0 & 0 & \cdots & {h\left( {{iM} + M - 1} \right)}\end{bmatrix}}$ $H = \begin{bmatrix}h_{0,0} & 0 & h_{1,0} & 0 \\0 & h_{0,1} & 0 & h_{1,1} \\0 & h_{0,0} & 0 & h_{1,0} \\h_{1,1} & 0 & h_{0,1} & 0\end{bmatrix}$

The construction of the matrix H and hi,0 and hi,1 may be described in aspecific example below. It may be assumed that the number of the users Nis equal to 2, the number of the sub-carriers M is equal to 8 and theroll-off factor r is equal to 0.5, and the coefficient h(n) of theprototype function may be obtained based on the RRC formula:

${{h(n)} = \frac{{\frac{4*0.5*n}{8}{\cos\left\lbrack \frac{{\pi\left( {1 + 0.5} \right)}n}{8} \right\rbrack}} + {\sin\left\lbrack \frac{{\pi\left( {1 - 0.5} \right)}n}{8} \right\rbrack}}{\left\lbrack {1 - \left( \frac{4*0.5*n}{8} \right)^{2}} \right\rbrack\pi\; n}},{{- 4} \leq n < 4},{n \neq 0},{- 4}$${h(0)} = {\frac{1}{8} + {\frac{0.5}{8}\left( {\frac{4}{\pi} - 1} \right)}}$${h\left( {- 4} \right)} = {{- \frac{0.5}{8}}\left\{ {{\frac{2}{\pi}{\cos\left\lbrack {\frac{\pi}{4*0.5}\left( {1 + 0.5} \right)} \right\rbrack}} - {\cos\left\lbrack {\frac{\pi}{4*0.5}\left( {1 - 0.5} \right)} \right\rbrack}} \right\}}$

The matrix element blocks h_(0,0),h_(0,1),h_(1,0),h_(1,1) may berespectively:

$h_{0,0} = \begin{bmatrix}{h(0)} & \; & \; & \; \\\; & {h(1)} & \; & \; \\\; & \; & {h(2)} & \; \\\; & \; & \; & {h(3)}\end{bmatrix}$ $h_{0,1} = \begin{bmatrix}{h(4)} & \; & \; & \; \\\; & {h(5)} & \; & \; \\\; & \; & {h(6)} & \; \\\; & \; & \; & {h(7)}\end{bmatrix}$ $h_{1,0} = \begin{bmatrix}{h(8)} & \; & \; & \; \\\; & {h(9)} & \; & \; \\\; & \; & {h(10)} & \; \\\; & \; & \; & {h(11)}\end{bmatrix}$ $h_{1,1} = \begin{bmatrix}{h(12)} & \; & \; & \; \\\; & {h(13)} & \; & \; \\\; & \; & {h(14)} & \; \\\; & \; & \; & {h(15)}\end{bmatrix}$

Finally, the obtained matrix H may be equal to

$H = \begin{bmatrix}h_{0,0} & 0 & h_{1,0} & 0 \\0 & h_{0,1} & 0 & h_{1,1} \\0 & h_{0,0} & 0 & h_{1,0} \\h_{1,1} & 0 & h_{0,1} & 0\end{bmatrix}$

A 2NM point symbol signal may be obtained by right-multiplying an NMpoint frequency domain signal X(k) by the coefficient matrix H, anM-point inverse fast Fourier transform (IFFT) repeated two timesoperation may be performed on the 2NM point symbol signal, and a 2NMpoint time domain signal may be obtained.

The receiving end may perform an inverse operation of the sending end.The receiving end may perform prefix removal processing on a receivedsignal, and then an equalizer may perform an equalization operation toremove channel interference. After the signal is transmitted to aninverse interleaving operation unit for processing, an M-point FFT unitmay perform an M-point FFT repeated 2N times on the signal to obtain afrequency domain sub-band signal. Then, a full-band signal may besynthesized via sub-band synthesizing and filtering, and transmitted toan NM-point IFFT unit to obtain a time domain signal. Symbol inverseordering may be performed on the signal output by the transform unit,and then a reconstructed symbol signal of the sending end may beobtained through processing of a symbol inverse mapping unit.

The present disclosure may adopt a frequency domain zero forcingequalizer. The frequency domain zero forcing equalizer may perform anFFT to transform the signal after prefix removal to frequency domain,and divide the signal by a system function H(k) (Fourier transform ofchannel impulse response h(n)), and finally perform an inverse fastFourier transform (IFFT) to obtain the signal with the channelinterference removed.

MC-TDMA may be a multi-carrier modulation system based on a filter bank,and use a modified DFT (MDFT) filter bank. MC-TDMA may be obtained byintroducing advantages of the single carrier modulation system based onthe conventional filter bank multi-carrier (FBMC) modulation system. TheMC-TDMA system may have double advantages of the multi-carrier andsingle carrier systems, and have the extremely low power peak to averagepower ratio (PAPR) and the strong anti-multipath fading capability in awireless channel and an anti-carrier asynchrony capability. The PAPRvalue may approach a theoretical minimum value, and may be lower thanthe PAPR value of the single carrier frequency division multiplexingaccess (SC-FDMA) system used in the LTE standard. The simulation mayshow that the anti-carrier asynchrony capability of the MC-TDMA systemis 10-fold or more of OFDM. The MC-TDMA modulation system may be usedfor both uplink communication and downlink communication. The MC-TDMAsystem may reduce power consumption of a terminal device and a basestation device, and simultaneity reduce a precision requirement for theterminal device clock frequency. The MC-TDMA system may have thefeatures of the flexible design and easy implementation, and may be usedfor both high-speed communication and Internet of things communication.

FIG. 3, FIG. 4 and FIG. 5 respectively show simulation results ofperformance comparison of the MC-TDMA modulation system provided by thepresent disclosure. FIG. 3 shows a simulation result of PAPR, in which srepresents symbol shift, a solid line represents the OFDMA system, abroken line represents the SC-FDMA system, and a dotted line representsthe MC-TDMA system. As best seen in FIG. 3, the PAPR value of theMC-TDMA system is the lowest, and has a very significant improvementcompared to OFDMA and SC-FDMA. FIG. 4 shows a comparison diagram of asymbol error rate, and it can be seen from the figure that the errorrate of MC-TDMA is lowest. FIG. 5 shows a comparison diagram of theperformance of the anti-sub-carrier drift in the systems, and it can beseen from the figure that in the case of carrier drift is 10%, MC-TDMAstill has excellent performance while OFDMA and SC-FDMA already cannotwork. It can be clearly seen from the simulation results that theperformance of the MC-TDMA system provided by the present disclosure isbetter than that of OFDMA and SC-FDMA. MC-TDMA can be used for bothuplink communication and downlink communication, and for both high-speedcommunication and asynchronous low-speed communication.

The embodiments recited in the present disclosure are as describedabove, but are merely illustrative of the present disclosure for thepurpose of understanding the present disclosure and are not intended tolimit the present disclosure. It may be apparent to those skilled in theart that various changes and modifications may be made in the form anddetails of the disclosure in accordance with the present disclosurewithout departing from the spirit and essence of the present disclosure,but the scope of patent protection of the present disclosure is subjectto the protection scope of the claims.

The invention claimed is:
 1. A modified discrete Fourier transform(MDFT) treatment used for multi-carrier time-division multiplexingmodulation, comprising: performing an inverse Fourier transform on afrequency domain symbol signal through an inverse Fourier transformmodule; and performing an interleaving operation after performing theinverse Fourier transform.
 2. The MDFT treatment of claim 1, whereinbefore performing the inverse Fourier transform, the MDFT treatmentfurther includes: performing a pre-filtering on a signal by a pre-filterto obtain the frequency domain symbol signal.
 3. The MDFT treatment ofclaim 2, wherein the pre-filter has symmetric coefficients.
 4. The MDFTtreatment of claim 2, wherein the frequency domain symbol signal is anNM point frequency domain symbol signal, and the performing thepre-filtering includes: performing a pre-filtering on the NM pointfrequency domain symbol signal; and generating a 2NM point frequencydomain symbol signal based on the frequency domain symbol signal and acoefficient matrix, wherein N and M are positive integers that arelarger than or equal to
 1. 5. The MDFT treatment of claim 4, wherein thegenerating the 2NM point frequency domain symbol signal based on the NMpoint frequency domain symbol signal and the coefficient matrixincludes: right-multiplying the NM point frequency domain symbol signalby the coefficient matrix.
 6. The MDFT treatment of claim 4, wherein thecoefficient matrix is determined based on a prototype filter function.7. The MDFT treatment of claim 6, wherein the coefficient matrixincludes a first sub-matrix and a second sub-matrix.
 8. The MDFTtreatment of claim 7, wherein the prototype filter function is a rootraised cosine (RRC) prototype function, and the first sub-matrix or thesecond sub-matrix is determined based on a coefficient function of theRRC prototype function.
 9. The MDFT treatment of claim 4, wherein theperforming the inverse Fourier transform includes: performing an M-pointIFFT repeated 2N times on the 2NM point frequency domain symbol signalto obtain a 2NM point time domain signal.
 10. The MDFT treatment ofclaim 9, wherein the performing the interleaving operation includes:performing the interleaving operation on the 2NM point time domainsignal to obtain an NM point complex symbol signal.
 11. An inversemodified discrete Fourier transform (IMDFT) treatment used formulti-carrier time-division multiplexing modulation, comprising:performing an inverse interleaving operation; and performing a Fouriertransform treatment on a signal to obtain a frequency domain symbolsignal through a Fourier transform module.
 12. The IMDFT treatment ofclaim 11, further comprising: performing a sub-band synthesizing andfiltering on the frequency domain symbol signal to obtain a full-bandfrequency domain signal.
 13. The IMDFT treatment of claim 12, whereinthe frequency domain symbol signal is a 2NM point frequency domainsymbol signal, and the performing the sub-band synthesizing andfiltering treatment includes: performing a post-filtering on the 2NMpoint frequency domain symbol signal; and generating an NM pointfrequency domain symbol signal based on the 2NM point frequency domainsymbol signal and a transpose matrix of a coefficient matrix, wherein Nand M are positive integers that are larger than or equal to
 1. 14. TheIMDFT treatment of claim 13, wherein the generating the NM pointfrequency domain symbol signal includes: right-multiplying the 2NM pointfrequency domain symbol signal by the transpose matrix of thecoefficient matrix.
 15. The IMDFT treatment of claim 13, wherein thecoefficient matrix is determined based on a prototype filter function.16. The IMDFT treatment of claim 13, wherein the performing the Fouriertransform treatment on the signal includes: performing an M-pointFourier transform repeated 2N times on the signal to obtain the 2NMpoint frequency domain symbol signal.
 17. An MDFT unit, comprising: aninverse Fourier transform module configured to perform an inverseFourier transform on a frequency domain symbol signal; and aninterleaving operation module configured to perform an interleavingoperation after performing the inverse Fourier transform.
 18. The MDFTunit of claim 17, further comprising: a sub-band analyzing and filteringmodule configured to perform a pre-filtering on a signal by a pre-filterbefore performing the inverse Fourier transform to obtain the frequencydomain symbol signal.